Method and system for channel estimation in a single channel MIMO system with multiple RF chains for WCDMA/HSDPA

ABSTRACT

Aspects of a method and system for channel estimation in a MIMO communication system with multiple RF chains for WCDMA/HSDPA may comprise receiving a plurality of communication signals from a plurality of transmit antennas. A plurality of vectors of baseband combined channel estimates may be generated based on phase rotation of the received plurality of communication signals. A matrix of processed baseband combined channel estimates may be generated based on the generated plurality of vectors of baseband combined channel estimates. A plurality of amplitude and phase correction signals may be generated based on the generated plurality of vectors of baseband combined channel estimates. An amplitude and a phase of at least a portion of the received plurality of communication signals may be adjusted based on the generated plurality of amplitude and phase correction signals, respectively.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This patent application is a continuation of U.S. Ser. No. 11/173,305filed Jun. 30, 2005, which application makes reference to, claimspriority to and claims benefit from U.S. provisional patent applicationSer. No. 60/616,687 filed Oct. 6, 2004.

This application also makes reference to:

U.S. application Ser. No. 11/173,870 filed Jun. 30, 2005;

U.S. application Ser. No. 11/174,303 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,502 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,871 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,964 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,252 filed Jun. 30, 2005;

U.S. application Ser. No. 11/174,252 filed Jun. 30, 2005;

U.S. application Ser. No. 11/172,756 filed Jun. 30, 2005;

U.S. application Ser. No. 11/172,759 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,689 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,304 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,129 filed Jun. 30, 2005;

U.S. application Ser. No. 11/172,779 filed Jun. 30, 2005;

U.S. application Ser. No. 11/172,702 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,727 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,726 filed Jun. 30, 2005;

U.S. application Ser. No. 11/172,781 filed Jun. 30, 2005;

U.S. application Ser. No. 11/174,067 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,854 filed Jun. 30, 2005;

U.S. application Ser. No. 11/173,911 filed Jun. 30, 2005; and

U.S. application Ser. No. 11/174,403 filed Jun. 30, 2005.

The above referenced applications are hereby incorporated herein byreference in their entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to channel estimation. Morespecifically, certain embodiments of the invention relate to a methodand system for channel estimation in a single channel multi-inputmulti-output (MIMO) system with multiple RF chains for WCDMA/HSDPA.

BACKGROUND OF THE INVENTION

Mobile communications has changed the way people communicate and mobilephones have been transformed from a luxury item to an essential part ofevery day life. The use of mobile phones is today dictated by socialsituations, rather than hampered by location or technology. While voiceconnections fulfill the basic need to communicate, and mobile voiceconnections continue to filter even further into the fabric of every daylife, the mobile Internet is the next step in the mobile communicationrevolution. The mobile Internet is poised to become a common source ofeveryday information, and easy, versatile mobile access to this datawill be taken for granted.

Third generation (3G) cellular networks have been specifically designedto fulfill these future demands of the mobile Internet. As theseservices grow in popularity and usage, factors such as cost efficientoptimization of network capacity and quality of service (QoS) willbecome even more essential to cellular operators than it is today. Thesefactors may be achieved with careful network planning and operation,improvements in transmission methods, and advances in receivertechniques. To this end, carriers need technologies that will allow themto increase downlink throughput and, in turn, offer advanced QoScapabilities and speeds that rival those delivered by cable modem and/orDSL service providers. In this regard, networks based on wideband CDMA(WCDMA) technology may make the delivery of data to end users a morefeasible option for today's wireless carriers.

FIG. 1A is a technology timeline indicating evolution of existing WCDMAspecification to provide increased downlink throughput. Referring toFIG. 1A, there is shown data rate spaces occupied by various wirelesstechnologies, including General Packet Radio Service (GPRS) 100,Enhanced Data rates for GSM (Global System for Mobile communications)Evolution (EDGE) 102, Universal Mobile Telecommunications System (UMTS)104, and High Speed Downlink Packet Access (HSDPA) 106.

The GPRS and EDGE technologies may be utilized for enhancing the datathroughput of present second generation (2G) systems such as GSM. TheGSM technology may support data rates of up to 14.4 kilobits per second(Kbps), while the GPRS technology, introduced in 2001, may support datarates of up to 115 Kbps by allowing up to 8 data time slots per timedivision multiple access (TDMA) frame. The GSM technology, by contrast,may allow one data time slot per TDMA frame. The EDGE technology,introduced in 2003, may support data rates of up to 384 Kbps. The EDGEtechnology may utilizes 8 phase shift keying (8-PSK) modulation forproviding higher data rates than those that may be achieved by GPRStechnology. The GPRS and EDGE technologies may be referred to as “2.5G”technologies.

The UMTS technology, introduced in 2003, with theoretical data rates ashigh as 2 Mbps, is an adaptation of the WCDMA 3G system by GSM. Onereason for the high data rates that may be achieved by UMTS technologystems from the 5 MHz WCDMA channel bandwidths versus the 200 KHz GSMchannel bandwidths. The HSDPA technology is an Internet protocol (IP)based service, oriented for data communications, which adapts WCDMA tosupport data transfer rates on the order of 10 megabits per second(Mbits/s). Developed by the 3G Partnership Project (3GPP) group, theHSDPA technology achieves higher data rates through a plurality ofmethods. For example, many transmission decisions may be made at thebase station level, which is much closer to the user equipment asopposed to being made at a mobile switching center or office. These mayinclude decisions about the scheduling of data to be transmitted, whendata is to be retransmitted, and assessments about the quality of thetransmission channel. The HSDPA technology may also utilize variablecoding rates. The HSDPA technology may also support 16-level quadratureamplitude modulation (16-QAM) over a high-speed downlink shared channel(HS-DSCH), which permits a plurality of users to share an air interfacechannel

In some instances, HSDPA may provide a two-fold improvement in networkcapacity as well as data speeds up to five times (over 10 Mbit/s) higherthan those in even the most advanced 3G networks. HSDPA may also shortenthe roundtrip time between network and terminal, while reducingvariances in downlink transmission delay. These performance advances maytranslate directly into improved network performance and highersubscriber satisfaction. Since HSDPA is an extension of the GSM family,it also builds directly on the economies of scale offered by the world'smost popular mobile technology. HSDPA may offer breakthrough advances inWCDMA network packet data capacity, enhanced spectral and radio accessnetworks (RAN) hardware efficiencies, and streamlined networkimplementations. Those improvements may directly translate into lowercost-per-bit, faster and more available services, and a network that ispositioned to compete more effectively in the data-centric markets ofthe future.

The capacity, quality and cost/performance advantages of HSDPA yieldmeasurable benefits for network operators, and, in turn, theirsubscribers. For operators, this backwards-compatible upgrade to currentWCDMA networks is a logical and cost-efficient next step in networkevolution. When deployed, HSDPA may co-exist on the same carrier as thecurrent WCDMA Release 99 services, allowing operators to introducegreater capacity and higher data speeds into existing WCDMA networks.Operators may leverage this solution to support a considerably highernumber of high data rate users on a single radio carrier. HSDPA makestrue mass-market mobile IP multimedia possible and will drive theconsumption of data-heavy services while at the same time reducing thecost-per-bit of service delivery, thus boosting both revenue andbottom-line network profits. For data-hungry mobile subscribers, theperformance advantages of HSDPA may translate into shorter serviceresponse times, less delay and faster perceived connections. Users mayalso download packet-data over HSDPA while conducting a simultaneousspeech call.

HSDPA may provide a number of significant performance improvements whencompared to previous or alternative technologies. For example, HSDPAextends the WCDMA bit rates up to 10 Mbps, achieving higher theoreticalpeak rates with higher-order modulation (16-QAM) and with adaptivecoding and modulation schemes. The maximum QPSK bit rate is 5.3 Mbit/sand 10.7 Mbit/s with 16-QAM. Theoretical bit rates of up to 14.4 Mbit/smay be achieved with no channel coding. The terminal capability classesrange from 900 kbit/s to 1.8 Mbit/s with QPSK modulation, and 3.6 Mbit/sand up with 16-QAM modulation. The highest capability class supports themaximum theoretical bit rate of 14.4 Mbit/s.

However, implementing advanced wireless technologies such as WCDMAand/or HSDPA may still require overcoming some architectural hurdles.For example, the RAKE receiver is the most commonly used receiver inCDMA systems, mainly due to its simplicity and reasonable performanceand WCDMA Release 99 networks are designed so that RAKE receivers may beused. A RAKE receiver contains a bank of spreading sequence correlators,each receiving an individual multipath. A RAKE receiver operates onmultiple discrete paths. The received multipath signals may be combinedin several ways, from which maximum ratio combining (MRC) is preferredin a coherent receiver. However, a RAKE receiver may be suboptimal inmany practical systems, for example, its performance may degrade frommultiple access interference (MAI), that is, interference induced byother users in the network.

In the case of a WCDMA downlink, MAI may result from inter-cell andintracell interference. The signals from neighboring base stationscompose intercell interference, which is characterized by scramblingcodes, channels and angles of arrivals different from the desired basestation signal. Spatial equalization may be utilized to suppressinter-cell interference. In a synchronous downlink application,employing orthogonal spreading codes, intra-cell interference may becaused by multipath propagation. Due to the non-zero cross-correlationbetween spreading sequences with arbitrary time shifts, there isinterference between propagation paths (or RAKE fingers) afterdespreading, causing MAI. The level of intra-cell interference dependsstrongly on the channel response. In nearly flat fading channels, thephysical channels remain almost completely orthogonal and intra-cellinterference does not have any significant impact on the receiverperformance. On the other hand, the performance of the RAKE receiver maybe severely deteriorated by intra-cell interference in frequencyselective channels. Frequency selectivity is common for the channels inWCDMA networks.

Due to the difficulties faced when non-linear channel equalizers areapplied to the WCDMA downlink, detection of the desired physical channelwith a non-linear equalizer may result in implementing an interferencecanceller or optimal multi-user receiver. Both types of receivers may beprohibitively complex for mobile terminals and may require informationnot readily available at the mobile terminal. Alternatively, the totalbase station signal may be considered as the desired signal. However,non-linear equalizers rely on prior knowledge of the constellation ofthe desired signal, and this information is not readily available at theWCDMA terminal. The constellation of the total base station signal, thatis, sum of all physical channels, is a high order quadrature amplitudemodulation (QAM) constellation with uneven spacing. The spacing of theconstellation changes constantly due to transmission power control (TPC)and possible power offsets between the control data fields,time-multiplexed to the dedicated physical channels. The constellationorder may also frequently change due to discontinuous transmission. Thismakes an accurate estimation of the constellation very difficult.

In this regard, the use of multiple transmit and/or receive antennas mayresult in an improved overall system performance. These multi-antennaconfigurations, also known as smart antenna techniques, may be utilizedto mitigate the negative effects of multipath and/or signal interferenceon signal reception. It is anticipated that smart antenna techniques maybe increasingly utilized both in connection with the deployment of basestation infrastructure and mobile subscriber units in cellular systemsto address the increasing capacity demands being placed on thosesystems. These demands arise, in part, from a shift underway fromcurrent voice-based services to next-generation wireless multimediaservices that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designedto introduce a diversity gain and to suppress interference generatedwithin the signal reception process. Such diversity gains improve systemperformance by increasing received signal-to-noise ratio, by providingmore robustness against signal interference, and/or by permittinggreater frequency reuse for higher capacity. In communication systemsthat incorporate multi-antenna receivers, a set of M receive antennasmay be utilized to null the effect of (M−1) interferers, for example.Accordingly, N signals may be simultaneously transmitted in the samebandwidth using N transmit antennas, with the transmitted signal thenbeing separated into N respective signals by way of a set of N antennasdeployed at the receiver. Systems that utilize multiple transmit andreceive antennas may be referred to as multiple-input multiple-output(MIMO) systems. One attractive aspect of multi-antenna systems, inparticular MIMO systems, is the significant increase in system capacitythat may be achieved by utilizing these transmission configurations. Fora fixed overall transmitted power, the capacity offered by a MIMOconfiguration may scale with the increased signal-to-noise ratio (SNR).For example, in the case of fading multipath channels, a MIMOconfiguration may increase system capacity by nearly M additionalbits/cycle for each 3-dB increase in SNR.

However, the widespread deployment of multi-antenna systems in wirelesscommunications, particularly in wireless handset devices, has beenlimited by the increased cost that results from increased size,complexity, and power consumption. Providing separate RF chain for eachtransmit and receive antenna is a direct factor that increases the costof multi-antenna systems. Each RF chain generally comprises a low noiseamplifier (LNA), a filter, a downconverter, and an analog-to-digitalconverter (A/D). In certain existing single-antenna wireless receivers,the single required RF chain may account for over 30% of the receiver'stotal cost. It is therefore apparent that as the number of transmit andreceive antennas increases, the system complexity, power consumption,and overall cost may increase. This poses problems for mobile systemdesigns and applications.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for channel estimation in a single channelmulti-input multi-output (MIMO) system with multiple RF chains forWCDMA/HSDPA, substantially as shown in and/or described in connectionwith at least one of the figures, as set forth more completely in theclaims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is a technology timeline indicating evolution of existing WCDMAspecification to provide increased downlink throughput.

FIG. 1B illustrates an exemplary HSDPA distributed architecture thatachieves low delay link adaptation, in connection with an embodiment ofthe invention.

FIG. 1C illustrates an exemplary Layer 1 HARQ control situated in a basestation to remove retransmission-related scheduling and storing from theradio network controller, in connection with an embodiment of theinvention.

FIG. 1D is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention.

FIG. 1E is a block diagram of exemplary 2 Tx antenna and M Rx antennawireless communication system with multiple RF chains and receiverchannel estimation, in accordance with an embodiment of the invention.

FIG. 2 illustrates an exemplary periodic phase rotation for an I signalcomponent, in accordance with an embodiment of the invention.

FIG. 3A is a block diagram of an exemplary system for providing phaserotation, channel estimation and for determining optimal phase andamplitude parameters or settings for an additional receive antenna, inaccordance with an embodiment of the invention.

FIG. 3B is a block diagram of an exemplary system for providing phaserotation, channel estimation and for determining optimal phase andamplitude parameters or setting for additional K−1 receive antennas, inaccordance with an embodiment of the invention.

FIG. 3C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention.

FIG. 4 is a block diagram of an exemplary baseband processor that may beutilized within a MIMO system, in accordance with an aspect of theinvention.

FIG. 5 is a block diagram of an exemplary system for determining channelestimation, in accordance with an embodiment of the invention.

FIG. 6 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized forchannel estimation in a 2-Tx and M-Rx antennas system, in accordancewith an embodiment of the invention.

FIG. 7 is a flowchart illustrating exemplary steps that may be utilizedfor channel estimation in a wireless communication system, in accordancewith an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method forchannel estimation in a communication system. Aspects of the method maycomprise receiving a plurality of communication signals from a pluralityof transmit antennas. A plurality of vectors of baseband combinedchannel estimates may be generated based on phase rotation of thereceived plurality of communication signals. A matrix of processedbaseband combined channel estimates may be generated based on thegenerated plurality of vectors of baseband combined channel estimates. Aplurality of amplitude and phase correction signals may be generatedbased on the generated plurality of vectors of baseband combined channelestimates. An amplitude and a phase of at least a portion of thereceived plurality of communication signals may be adjusted based on thegenerated plurality of amplitude and phase correction signals,respectively.

A plurality of weights may be determined that may be applied to each ofthe received plurality of communication signals based on the generatedplurality of amplitude and phase correction signals. The phase andamplitude of at least a portion of the received plurality ofcommunication signals may be adjusted based on the determined pluralityof weights. The determined plurality of weights may be calculatedutilizing an adaptive algorithm such as a least mean squares algorithm,for example. The received plurality of communication signals may befurther filtered, amplified and converted to digital signals. Thegenerated plurality of baseband combined channel estimates may bederived via rotation at additional antennas. The received plurality ofcommunication signals may be modulated into in phase (I) components andquadrature (Q) components.

FIG. 1B illustrates an exemplary HSDPA distributed architecture thatachieves low delay link adaptation, in connection with an embodiment ofthe invention. Referring to FIG. 1B, there is shown terminals 110 and112 and a base station (BS) 114. HSDPA is built on a distributedarchitecture that achieves low delay link adaptation by placing keyprocessing at the BS 114 and thus closer to the air interface asillustrated. Accordingly, the MAC layer at the BS 114 is moved fromLayer 2 to Layer 1, which implies that the systems may respond in a muchfaster manner with data access. Fast link adaptation methods, which aregenerally well established within existing GSM/EDGE standards, includefast physical layer (L1) retransmission combining and link adaptationtechniques. These techniques may deliver significantly improved packetdata throughput performance between the mobile terminals 110 and 112 andthe BS 114.

The HSDPA technology employs several important new technologicaladvances. Some of these may comprise scheduling for the downlink packetdata operation at the BS 114, higher order modulation, adaptivemodulation and coding, hybrid automatic repeat request (HARQ), physicallayer feedback of the instantaneous channel condition, and a newtransport channel type known as high-speed downlink shared channel(HS-DSCH) that allows several users to share the air interface channel.When deployed, HSDPA may co-exist on the same carrier as the currentWCDMA and UMTS services, allowing operators to introduce greatercapacity and higher data speeds into existing WCDMA networks. HSDPAreplaces the basic features of WCDMA, such as variable spreading factorand fast power control, with adaptive modulation and coding, extensivemulticode operation, and fast and spectrally efficient retransmissionstrategies.

In current-generation WCDMA networks, power control dynamics are on theorder of 20 dB in the downlink and 70 dB in the uplink. WCDMA downlinkpower control dynamics are limited by potential interference betweenusers on parallel code channels and by the nature of WCDMA base stationimplementations. For WCDMA users close to the base station, powercontrol may not reduce power optimally, and reducing power beyond the 20dB may therefore have only a marginal impact on capacity. HSDPA, forexample, utilizes advanced link adaptation and adaptive modulation andcoding (AMC) to ensure all users enjoy the highest possible data rate.AMC therefore adapts the modulation scheme and coding to the quality ofthe appropriate radio link.

FIG. 1C illustrates an exemplary Layer 1 HARQ control situated in a basestation to remove retransmission-related scheduling and storing from theradio network controller, in connection with an embodiment of theinvention. Referring to FIG. 1C, there is shown a hybrid automaticrepeat request (HARQ) operation, which is an operation designed toreduce the delay and increase the efficiency of retransmissions. Layer 1HARQ control is situated in the Node B, or base station (BS) 174, thusremoving retransmission-related scheduling and storing from the radionetwork controller RNC 172. This HARQ approach avoids hub delay andmeasurably reduces the resulting retransmission delay.

For example, when a link error occurs, due to signal interference orother causes, a mobile terminal 176 may request the retransmission ofthe data packets. While current-generation WCDMA networks handle thoseretransmission requests through the radio network controller 172, HSDPAretransmission requests are managed at the base station 174. Using thisapproach, packets are combined at the physical (PHY) layer and themobile terminal 176 stores the received data packets in soft memory. Ifdecoding has failed, the new transmission is combined with the oldtransmission before channel decoding. The HSDPA approach allowspreviously transmitted bits from the original transmission to becombined with the retransmission. This combining strategy providesimproved decoding efficiencies and diversity gains while minimizing theneed for additional repeat requests. The combining may include differentpuncturing pattern therefore providing code and time diversity.

While the spreading factor may be fixed, the coding rate may varybetween ¼ and ¾, and the HSDPA specification supports the use of 5, 10or 15 multicodes, for example. More robust coding, fast HARQ, andmulti-code operation eliminates the need for variable spreading factorand also allows for more advanced receiver structures in the mobile suchas equalizers as apposed to the traditional RAKE receiver used in mostCDMA systems. This approach may also allow users having good signalquality or higher coding rates and those at the more distant edge of thecell having lower coding rates to each receive an optimum available datarate.

By moving data traffic scheduling to the base station 174, and thuscloser to the air interface, and by using information about channelquality, terminal capabilities, QoS, and power/code availability, HSDPAmay achieve more efficient scheduling of data packet transmissions.Moving these intelligent network operations to the base station 174allows the system to take full advantage of short-term variations, andthus to speed and simplify the critical transmission scheduling process.The HSDPA approach may, for example, manage scheduling to track the fastfading of the users and when conditions are favorable to allocate mostof the cell capacity to a single user for a very short period of time.At the base station 174, HSDPA gathers and utilizes estimates of thechannel quality of each active user. This feedback provides currentinformation on a wide range of channel physical layer conditions,including power control, ACK/NACK ratio, QoS, and HSDPA-specific userfeedback.

While WCDMA Release 99 or WCDMA Release 4 may support a downlink channel(DCH) or a downlink shared channel (DSCH), the HSDPA operation providedby WCDMA Release 5 may be carried on a high-speed downlink sharedchannel (HS-DSCH). This higher-speed approach uses a 2 ms frame length,compared to DSCH frame lengths of 10, 20, 40 or 80 ms. DSCH utilizes avariable spreading factor of 4 to 256 chips while HS-DSCH may utilize afixed spreading factor of 16 with a maximum of 15 codes. HS-DSCH maysupports 16-level quadrature amplitude modulation (16-QAM), linkadaptation, and the combining of retransmissions at the physical layerwith HARQ. HSDPA also leverages a high-speed shared control channel(HS-SCCH) to carry the required modulation and retransmissioninformation. An uplink high-speed dedicated physical control channel(HS-DPCCH) may carry ARQ acknowledgements, downlink quality feedback andother necessary control information on the uplink.

FIG. 1D is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention. Referring to FIG. 1D, in practicaldeployments, HSDPA more than doubles the achievable peak user bit ratescompared to WCDMA Release 99. With bit rates that are comparable to DSLmodem rates, HS-DSCH may deliver user bit rates in large macrocellenvironments exceeding 1 Mbit/s, and rates in small microcells up to 5Mbit/s. The HSDPA approach supports both non-real-time UMTS QoS classesand real-time UMTS QoS classes with guaranteed bit rates.

Cell throughput, defined as the total number of bits per secondtransmitted to users through a single cell, increases 100% with HSDPAwhen compared to the WCDMA Release 99. This is because HSDPA's use ofHARQ combines packet retransmission with the earlier transmission, andthus no transmissions are wasted. Higher order modulation schemes, suchas 16-QAM, enable higher bit rates than QPSK-only modulation in WCDMARelease 99, even when the same orthogonal codes are used in bothsystems. The highest throughput may be obtained with low inter-pathinterference and low inter-cell interference conditions. In microcelldesigns, for example, the HS-DSCH may support up to 5 Mbit/s per sectorper carrier, or 1 bit's/Hz/cell.

FIG. 1E is a block diagram of exemplary 2 Tx antenna and M Rx antennawireless communication system with multiple RF chains and receiverchannel estimation, in accordance with an embodiment of the invention.Referring to FIG. 1E, the wireless system 100 may comprise a dedicatedphysical channel (DPCH) block 126, a plurality of mixers 128, 130 and132, a plurality of combiners 134 and 136, a first transmit antenna (Tx1) 138 and an additional transmit antenna (Tx 2) 140 on the transmitside. On the receive side, the wireless system 100 may comprise aplurality of receive antennas 106 _(1 . . . M), a single weightgenerator (SWG) 110, a plurality of RF blocks 114 _(1 . . . P), aplurality of chip matched filters (CMF) 116 _(1 . . . P), a baseband(BB) processor 126 and a single weight generator baseband processor(SWGBB) 121. The SWGBB 121 may comprise a channel estimator 122 and asingle weight generator (SWG) algorithm block 124.

The DPCH 126 may be adapted to receive a plurality of input channels,for example, a dedicated physical control channel (DPCCH) and adedicated physical data channel (DPDCH). The DPCH 126 may simultaneouslycontrol the power of DPCCH and DPDCH. The mixer 128 may be adapted tomix the output of DPCH 126 with a spread and/or scrambled signal togenerate a spread complex valued signal that may be input to mixers 130and 132. The mixers 130 and 132 may weight the complex valued inputsignals with weight factors W₁ and W₂, respectively, and may generateoutputs to a plurality of combiners 134 and 136 respectively. Thecombiners 134 and 136 may combine the outputs generated by mixers 130and 132 with common pilot channel 1 (CPICH1) and common pilot channel 2(CPICH2) respectively. The common pilot channels 1 and 2 may have afixed channelization code allocation that may be utilized to measure thephase amplitude signal strength of the channels. The weights W₁ and W₂may be utilized, for example, phase and or amplitude adjustments and maybe generated by the single weight generator (SWG) algorithm block 124.The antennas 138 and 140 may receive the generated outputs from thecombiners 134 and 136 and may transmit wireless signals.

The plurality of receive antennas 106 _(1 . . . M) may each receive atleast a portion of the transmitted signal. The SWG 110 may comprisesuitable logic, circuitry, and/or code that may be adapted to determinea plurality of weights to be applied to each of the input signalsR_(1 . . . M). The SWG 110 may be adapted to modify the phase andamplitude of a portion of the transmitted signals received by theplurality of receive antennas 106 _(1 . . . M) and generate a pluralityof output signals RF_(1 . . . P).

The plurality of RF blocks 114 _(1 . . . P) may comprise suitable logic,circuitry, and/or code that may be adapted to process an RF signal. TheRF blocks 114 _(1 . . . P) may perform, for example, filtering,amplification, and analog-to-digital (A/D) conversion operations. Theplurality of transmit antennas 138 and 140 may transmit the processed RFsignals to a plurality of receive antennas 106 _(1 . . . M). The singleweight generator SWG 110 may comprise suitable logic, circuitry, and/orcode that may be adapted to determine a plurality of weights, which maybe applied to each of the input signals. The single weight generator SWG110 may be adapted to modify the phase and amplitude of at least aportion of the signals received by the plurality of receive antennas 106_(1 . . . M) and generate a plurality of output signals RF_(1 . . . P).The plurality of RF receive blocks 114 _(1 . . . P) may comprisesuitable logic, circuitry and/or code that may be adapted to amplify andconvert the received analog RF signals RF_(1 . . . P) down to baseband.The plurality of RF receive blocks 114 _(1 . . . P) may each comprise ananalog-to-digital (A/D) converter that may be utilized to digitize thereceived analog baseband signal.

The plurality of chip matched filters (CMF) 116 _(1 . . . P) maycomprise suitable logic, circuitry and/or code that may be adapted tofilter the output of the plurality of RF receive blocks 114 _(1 . . . P)so as to produce in-phase (I) and quadrature (Q) components (I, Q). Inthis regard, in an embodiment of the invention, the plurality of chipmatched filters (CMF) 116 _(1 . . . P) may comprise a pair of digitalfilters that are adapted to filter the I and Q components to within thebandwidth of WCDMA baseband (3.84 MHz). The outputs of the plurality ofchip matched filters (CMF) 116 _(1 . . . P) may be transferred to the BBprocessor 126.

The BB 126 may be adapted to receive a plurality of in-phase andquadrature components (I, Q) from a plurality of chip matched filters(CMF) 116 _(1 . . . P) and generate a plurality of baseband combinedchannel estimates {circumflex over (h)}₁ to {circumflex over (h)}_(P).The BB 126 may be adapted to generate a plurality of estimates{circumflex over (X)}₁ to {circumflex over (X)}_(P) of the originalinput spatial multiplexing sub-stream signals or symbols X₁ to X_(P).The BB 126 may be adapted to separate the different space-time channelsutilizing a Bell Labs Layered Space-Time (BLAST) algorithm, for example,by performing sub-stream detection and sub-stream cancellation. Thecapacity of transmission may be increased almost linearly by utilizingthe BLAST algorithm.

The plurality of cluster path processors CPP 118 _(1 . . . P) maygenerate a plurality of baseband combined channel estimates {circumflexover (h)}₁ to {circumflex over (h)}_(P) that may correspond to theplurality of receive antennas 106 _(1 . . . M). The channel estimator122 may comprise suitable logic, circuitry, and/or code that may beadapted to process the received estimates {circumflex over (h)}₁ to{circumflex over (h)}_(P) from the BB processor 126 and may generate amatrix Ĥ of processed estimated channels that may be utilized by thesingle weight generator (SWG) algorithm block 124.

The SWG algorithm block 124 may determine a plurality of amplitude andphase values A_(i) and φ_(I), respectively, which may be utilized by SWG110 to modify the phase and amplitude of a portion of the transmittedsignals received by the plurality of receive antennas 106 _(1 . . . M)and generate a plurality of output signals RF_(1 . . . P).

FIG. 2 illustrates an exemplary periodic phase rotation for an I signalcomponent, in accordance with an embodiment of the invention. Referringto FIG. 2, for the wireless system 200 in FIG. 1, by rotating the phaseat the receive antennas 206 _(1 . . . M) from 0 to 360 degrees, it maybe possible to estimate all propagation channels, h_(1 . . . M), at thesame time utilizing complex multiplication and integration. Thisoperation is equivalent to making all the channels at the Rx antennasorthogonal. FIG. 2 illustrates the periodic rotation of the I componentin an RF signal.

FIG. 3A is a block diagram of an exemplary system for providing phaserotation, channel estimation and for determining optimal phase andamplitude parameters or settings for an additional receive antenna, inaccordance with an embodiment of the invention. Referring to FIG. 3A, areceiver system 300 may comprise a first receive antenna Rx 1 302, anadditional antenna Rx 2 304, a combiner 306, a complex multiplier 308,and a single weight generator baseband (SWGBB) processor 310. The SWGBBprocessor 310 may comprise a phase rotation start controller block 314,a delay block 316, a SWG channel estimator 318, a single weightgenerator (SWG) algorithm block 320, and a RF phase and amplitudecontroller 312. The SWGBB processor 310 provides similar functionalityas the SMBB processor 126 in FIG. 1.

The receive antennas Rx 1 302 and Rx 2 304 may each receive a portion ofthe transmitted signal. The combiner 306 may be adapted to combine thereceived signals into a single RF signal RF₁, for example. The complexmultiplier 308 may be adapted to receive a plurality of input signalsfrom the additional receive antenna Rx 2 304 and the RF phase andamplitude controller 312 and may generate an output signal to thecombiner 306.

The phase rotation start controller block 314 may comprise suitablelogic, circuitry and/or that may be adapted to start after receiving areset signal and may generate a plurality of output signals to the delayblock 316 and the RF phase and amplitude controller 312. The delay block316 may be adapted to receive an input signal from the phase rotationstart controller block 314 and generate a delayed output signal to theSWG channel estimator 318. The SWG channel estimator 318 may comprisesuitable logic, circuitry, and/or code that may be adapted to processthe received baseband combined channel estimates per transmit antenna ĥ₁. . . ĥ_(N) from the SMBB processor 126 and may generate a matrixĤ_(2×N) of processed estimated channels. The SWG channel estimator 318may be adapted to generate an algorithm start signal indicating the endof integration that may be utilized by the single weight generator (SWG)algorithm block 320.

The SWG algorithm block 320 may be adapted to receive a plurality ofsignals from the SWG channel estimator 318, for example, a matrixĤ_(2×N) of processed baseband combined channel estimates, an algorithmstart signal from the SWG channel estimator 318 and a noise powerestimation signal. The SWG algorithm block 320 may generate phase andamplitude correction signals and an algorithm end signal to the RF phaseand amplitude controller 312. The RF phase and amplitude controller 312may be adapted to receive the phase and amplitude values and thealgorithm end signal to modify the phase and amplitude of a portion ofthe transmitted signals received by the receive antenna Rx 2 302 andgenerate an output signal RF₁.

The SWG channel estimator 318 may receive baseband combined channelestimates ĥ₁ . . . ĥ_(N), which may include all transmission channelsfrom N Tx antennas and each Tx antenna may have a different channelestimation sequence, so that the different combined channels ĥ₁ . . .ĥ_(N) may be separated and estimated. The SWG channel estimator 318 maygenerate a matrix of channel estimates Ĥ_(2×N) to the SWG algorithmblock 320. A reset signal may be utilized to start phase rotation block314. The combined channel estimates from the BB 126 in FIG. 1 may betransferred to the channel estimator 318 for processing. When processingis complete, the SWG channel estimator 318 may indicate to the SWGalgorithm block 320 that the determination of the appropriate phase andamplitude correction for the portion of the received signal in theadditional antenna Rx 2 304 may start. The SWG algorithm block 320 mayutilize an estimation of the noise power and interference in determiningthe phase and amplitude values in addition to the matrix of channelestimates Ĥ_(2×N). The SWG algorithm block 320 may indicate to the RFphase and amplitude controller 312 the end of the weight determinationoperation and may then transfer to the RF phase and amplitude controller312, the determined phase and amplitude values. The RF phase andamplitude controller 312 may then modify the portion of the receivedsignal in the additional antenna Rx 2 304 via the complex multiplier308.

In operation, the RF phase and amplitude controller 312 may apply thesignal e^(jw) ^(r) ^(t) to the mixer 308 in FIG. 3A based on controlinformation provided by the phase rotator start controller 314. Theswitch 340 may select the rotation waveform source 342 based on thecontrol information provided by the phase rotator start controller 314.Once the channel weights are determined by the SWG algorithm block 320and the phase and amplitude components have been transferred to the RFphase and amplitude controller 312, the algorithm end signal may beutilized to change the selection of the switch 340. In this regard, theswitch 340 may be utilized to select and apply the signal Ae^(jφ) to themixer 308 in FIG. 3A.

FIG. 3B is a block diagram of an exemplary system for providing phaserotation, channel estimation and for determining optimal phase andamplitude parameters or setting for additional K−1 receive antennas, inaccordance with an embodiment of the invention. Referring to FIG. 3B, areceiver system 330 may correspond to a portion of the wirelesscommunication system 100 in FIG. 1 and may differ from the receiversystem 300 in FIG. 3A in that (K−1) additional receive antennas, Rx_2304 to Rx_K 305, and (K−1) mixers 308 to 309 may be utilized. Thecombiner 306 may combine the received signals into a single RF signalRF₁, for example. In this regard, the SWG channel estimator 318 may beadapted to process the combined channel estimates, ĥ₁ . . . ĥ_(N), anddetermine the propagation channel matrix estimate Ĥ_(K×N).

Referring to the FIG. 1, multiple receive antennas may be connected toeach of the RF chains RF₁ . . . RF_(N) as shown in FIG. 3B for thesingle RF chain RF₁. In this regard, the combined channel estimates ĥ₁ .. . ĥ_(N) and consequently the channel estimate matrix Ĥ_(K×N) may bedetermined per each RF chain RF₁ . . . RF_(N). Consequently, followingthis example, N matrices Ĥ_(K×N) may form a channel estimate matrixĤ_(M×N) in FIG. 1 (M=NK).

The SWG algorithm block 320 may also be adapted to determine (K−1)channel weights per RF chain, that may be utilized to maximize receiverSINR, for example, to be applied to the mixers 308 to 309 to modify theportions of the transmitted single channel communication signalsreceived by the additional receive antennas Rx_2 304 to Rx_K 305. The(K−1) channel weights per RF chain may comprise amplitude and phasecomponents, A₁ to A_(K−1) and φ₁ to φ_(K−1). The RF phase and amplitudecontroller 312 may also be adapted to apply rotation waveforms e^(jw)^(r1) ^(t) to e^(jw) ^(r(K−1)) ^(t) or phase and amplitude components,A₁ to A_(K−1) and φ₁ to φ_(K−1), to the mixers 308 to 309. In thisregard, the RF phase and amplitude controller 312 may apply the rotationwaveforms or the amplitude and phase components in accordance with thecontrol signals provided by the phase rotator start controller 314and/or the algorithm end signal generated by the SWG algorithm block320.

FIG. 3C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention. Referringto FIG. 3C, the RF phase and amplitude controller 312 may comprise aswitch 340, rotation waveform sources 342, and a plurality of SWGalgorithm determined weights 344. The switch 340 may comprise suitablehardware, logic, and/or circuitry that may be adapted to select betweenthe rotation waveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(K−1)) ^(t) andthe SWG algorithm determined weights A₁e^(jφ) ¹ to A_(K−1)e^(jφ) ^(K−1). The rotation waveform source 342 may comprise suitable hardware, logicand/or circuitry that may be adapted to generate the signal e^(jw) ^(rk)^(t), where w_(rk)=2πf_(rk) and f_(rk) is the rotation frequency thatpreserves orthogonality of the received signals at the multiplereceiving antennas. The rotation frequency that preserves the signalorthogonality at the receiving antennas may be selected as w_(rk)=kw_(r)where k=1, 2, 3 . . . K−1. Other rotation waveforms such as triangularor square may be utilized with the same frequency relationships. Inaddition, waveforms representing different orthogonal codes of the samefrequency may be utilized, similar to the CDMA orthogonal codes with thesame spreading. In this embodiment e^(jw) ^(rk) ^(t) is used as anexemplary waveform. The weights 344 may comprise suitable hardware,logic, and/or circuitry that may be adapted to generate the signalsA₁e^(jφ) ¹ to A_(K−1)e^(jφ) ^(K−1) from the amplitude and phasecomponents, A₁ to A_(K−1) and φ₁ to φ_(K−1), respectively.

In operation, the RF phase and amplitude controller 312 may apply thesignals e^(jw) ^(r1) ^(t) to e^(jw) ^(r(K−1)) ^(t) to the mixers 308 to309 in FIG. 3B based on control information provided by the phaserotator start controller 314. The switch 340 may select the rotationwaveform source 342 based on the control information provided by thephase rotator start controller 314. Once the channel weights aredetermined by the SWG algorithm block 320 and the phase and amplitudecomponents have been transferred to the RF phase and amplitudecontroller 312, the algorithm end signal may be utilized to change theselection of the switch 340. In this regard, the switch 340 may beutilized to select and apply the signals A₁e^(jφ) ¹ to A^(K−1)e^(jφ)^(M−1) to the mixers 308 to 309 in FIG. 3B.

FIG. 4 is a block diagram of an exemplary baseband processor that may beutilized within a MIMO system, in accordance with an aspect of theinvention. Referring to FIG. 4, the baseband processor 400 may comprisea cluster path processor (CPP) block 432, a maximum ratio combining(MRC) block 424, a despreader block 426, a diversity processor block428, a macrocell combiner block 430, a bit rate processing block 431, aconvolutional decoder block 438, and a turbo decoder block 440.

U.S. application Ser. No. 11/173,854 provides a detailed description ofsignal clusters and is hereby incorporated herein by reference in itsentirety.

The CPP block 432 may comprise a plurality of cluster processors thatmay be adapted to receive and process an input signal 502 received froma chip matched filter (CMF), for example. In the baseband receiverprocessor 400, the CPPs 432 a, . . . , 432 n within the CPP block 432may be partitioned into pairs of processors, wherein each pair ofprocessor may be adapted to track time-wise and estimate the complexphase and amplitude of the element in the cluster. A cluster maycomprise an aggregate of received multipath signals with maximum (max)time difference that may be no more than 16×1/3.84e6 seconds, forexample. Under these circumstances, the need for two processors may bederived from the fact that the WCDMA standard facilitates a receivingmode in which the transmitted signal is transmitted over two antennas,which necessitates the two processors. These receiving modes compriseclose loop 1 (CL1), close loop 2 (CL2), and STTD. The CPP block 432 maybe adapted to determine estimates of the entire transfer function of thechannel and may recover channels on a per base station basis.

The CPP block 432 may be adapted to generate channel estimates ĥ₁ and ĥ₂of the actual time varying impulse response of the channel per basestation. The CPP 432 may also generate timing information T on per basestation basis related to signals received by antennas at the receiveside, such as antennas 106 _(1 . . . M) of FIG. 1E, for example.Corresponding lock indicators L₁ and L₂ may also be generated by thecluster processors. The lock indicators may provide an indication ofwhich components in the corresponding estimates comprise valid componentvalues. In one embodiment of the invention, cluster path processors 432a, . . . , 432 n may be configured to operate in pairs when atransmitted signal is transmitted by two antenna, where the two antennamay be located in the same base station, or at different base stations.The channel estimates ĥ₁ and ĥ₂ of the actual time varying impulseresponse of the channel per base station, as well as lock indicators L1and L2, and the timing information T per base station may becommunicated to a single weight generation (SWG) block, for example, aswell as to the maximum-ratio combining (MRC) block 424 for furtherprocessing. The channel estimates ĥ₁ and ĥ₂, the lock indicators L1 andL2, and the timing information T may be utilized by an SWG block forgenerating a single weight (SW) control signal for phase shifting of oneor more signals received by receiver antennas.

The maximum-ratio combining block 424 may comprise suitable logic,circuitry and/or code to receive timing reference signals, T, andchannel estimates and lock indicators, (ĥ1, L1) and (ĥ2, L2), from thecorresponding cluster path processor block 432, which may be utilized bythe maximum-ratio combining block 424 to process received signals from achip matched filter (CMF) block, for example. The maximum ratiocombining block 424 may utilize channel estimate components that arevalid in accordance with the corresponding lock indicator. Channelestimate components that are not valid, in accordance with thecorresponding lock indicator, may not be utilized. The maximum-ratiocombining block 424 may be adapted to provide a combining scheme ormechanism for implementing a rake receiver which may be utilized withadaptive antenna arrays to combat noise, fading, and/or co-channelinterference.

In accordance with an embodiment of the invention, the maximum-ratiocombining block 424 may comprise suitable logic, circuitry, and/or codethat may be adapted to add individual distinct path signals, receivedfrom the assigned RF channel, together in such a manner to achieve thehighest attainable signal to noise ratio (SNR). The highest attainableSNR may be based upon a maximum ratio combiner. A maximum ratio combineris a diversity combiner in which each of multipath signals from allreceived multipaths are added together, each with unique gain. The gainof each multipath before summing can be made proportional to receivedsignal level for the multipath, and inversely proportional to themultipath noise level. Each of the maximum-ratio combining blocks may bealso adapted to utilize other techniques for signal combining suchselection combiner, switched diversity combiner, equal gain combiner, oroptimal combiner.

In one embodiment of the invention, the assignment of fingers in themaximum-ratio combining block 424 may be based on channel estimates h1and h2 from the cluster path processor block 432. The proportionalityconstants utilized in the maximum-ratio combining block 424 may be basedon the valid channel estimates, ĥ1 and ĥ2, from the cluster pathprocessor block 432.

The despreader (DS) block 426 may comprise a plurality of despreaderblocks 426 a, . . . , 426 n. Each of the despreader blocks 426 a, . . ., 426 n may comprise suitable logic, circuitry, and/or code that may beadapted to despread received signals that may have been previouslyspread through the application of orthogonal spreading codes in thetransmitter. Prior to transmission of an information signal, known as a“symbol”, the transmitter may have applied an orthogonal spreading codethat produced a signal Comprising a plurality of chips. The DS block 426may be adapted to generate local codes, for example Gold codes ororthogonal variable spreading factor (OVSF) codes, that may be appliedto received signals through a method that may comprise multiplicationand accumulation operations. Processing gain may be realized aftercompletion of integration over a pre-determined number of chips in whichthe symbol is modulated.

Following despreading at the receiver, the original symbol may beextracted. WCDMA may support the simultaneous transmission of aplurality of spread spectrum signals in a single RF signal by utilizingspreading codes among the spread spectrum signals which are orthogonalto reduce multiple access interference (MAI). The receiver may extractan individual symbol from the transmitted plurality of spread spectrumsignals by applying a despreading code, which may be equivalent to thecode that was utilized for generating the spread spectrum signal.Similarly to the CPP block 432 and the MRC block 424, the DS block 426may be assigned on a per base station basis, with the MRC block 424communicating with the DS block 426 that may be assigned to the samebase stations.

The diversity processor 428, comprising a plurality of diversityprocessor blocks 428 a, . . . , 428 n, may comprise suitable logic,circuitry, and/or code that may be adapted to combine signalstransmitted from multiple antennas in diversity modes. The diversitymodes may comprise OL, CL1 and CL2. The diversity processor 428 maycombine signals transmitted from multiple antennas that are located atthe same base station. Similarly with the cluster path processors 432,the maximum-ratio combining blocks 424, and the despreader blocks 426,the diversity processors 428 may be assigned on a per base stationbasis, with the diversity processors 428 communicating with despreaderblocks 426 that may be assigned to the same base stations.

The macrocell combiner 430 may comprise suitable logic, circuit and/orcode and may be adapted to achieve macroscopic diversity. Themacroscopic diversity scheme may be utilized for combining two or morelong-term lognormal signals, which may be obtained via independentlyfading paths received from two or more different antennas at differentbase-station sites. The microscopic diversity schemes may be utilizedfor combining two or more short-term Rayleigh signals, which areobtained via independently fading paths received from two or moredifferent antennas but only one receiving site.

The bit rate processing block 431 may comprise suitable logic, circuitryand/or code to process frames of data received from the macrocellcombiner 430. The processing may further comprise depuncturing, anddeinterleaving data in the received frame, and further determining arate at which processed frames are communicated in output signals.

The convolutional decoder 438 may comprise suitable logic, circuitryand/or code that may be utilized to handle decoding of convolutionalcodes as indicated in the 3GPP specification. The output of theconvolutional decoder may be a digital signal, which comprises voiceinformation, suitable for processing by a voice-processing unit. Theturbo decoder 440 may comprise suitable logic, circuitry and/or codethat may be utilized to handle decoding of turbo codes as indicated inthe 3GPP specification. The output of the turbo decoder 440 may be adigital signal, which has data information, such that it may be suitablefor use by a video display processor.

The maximum-ratio combining block 424 may be adapted to utilize thechannel estimates and lock indicators (ĥ1, L1), (ĥ2, L2) and timinginformation T per base station to assign rake fingers to receivedindividual distinct path signals and to assign proportionality constantsto each finger. Received individual distinct path signals may beprocessed in the maximum-ratio combining block 424 as signal clusterscomprising a plurality of received individual distinct path signals. Inan embodiment of the invention, the maximum-ratio combining block 424may assign a time, T(n), to the nth grid element of the CPP 432, wherethe plurality of times T(n) may be based on the timing reference T.Given a time assignment, and a time offset, toff, a given CPP 432, n,may detect an individual distinct path signal that is received during atime interval starting at [T(n)−toff/2], and ending at [T(n)+toff/2].

The individual distinct path signals received collectively for each CPP432 may constitute a signal cluster. The relationship of the values T(n)among the processing elements of the CPP 432 in the receiver may be suchthat T(n+1)−T(n) is equal to a constant value for values of n among theset of fingers. Thus, once T is determined, the timing relationships forthe receipt of the plurality of individual distinct path signalsconstituent in the signal cluster may be determined. The time offsetvalue, toff, may represent a time duration, which is at least as long asthe period of time required for transmitting the plurality of chipscontained in a symbol. For example, if the symbol comprises 16 chips,and the W-CDMA chip rate is 3.84×106 chips/second, then the time offsettoff may be (16/3.84×106) seconds or approximately 4 microseconds.

Embodiments of the invention may not be limited to values of thedifference T(n+1)−T(n) being constant among all n fingers in a rakereceiver. However, each value, T(n), may be based on the timingreference signal, T.

The maximum-ratio combining block 424 may proportionately scale and addthe received individual distinct path signals to produce a chip leveloutput, which may be communicated to the despreader block 426. Thedespreader block 426 may be adapted to despread the chip level signalreceived from the maximum-ratio combining block 424 to generateestimates of the original transmitted signals. The diversity processorblock 428 may be adapted to provide diversity processing and to generateoutput data estimates on a per base station basis. The macrocellcombiner block 430 may achieve macroscopic diversity when a receivedsignal has been transmitted by a plurality of base stations. The bitrate processing block 431 may perform processing tasks comprisingdepuncture and deinterleave on received frames of data that arecommunicated in received individual distinct path signals. The bit rateprocessing block 431 may determine a rate at which to communicateprocessed frames of data to the convolutional decoder block 438, and/orthe turbo decoder block 440. The convolution decoder block 438 may beadapted to perform convolutional decoding on the voice portion of thesignal generated from an output of the bit rate processing block 431.The turbo decoder block 440 may be adapted to perform turbo decoding onthe data portion of the signal generated from an output of the bit rateprocessing block 431.

FIG. 5 is a block diagram of an exemplary system for determining channelestimation, in accordance with an embodiment of the invention. Referringto FIG. 5, the SWG channel estimator 500 may comprise a phase rotator502, a complex combiner 506, a first integrator 504 and a secondintegrator 508.

The SWG channel estimator 500 may comprise suitable logic, circuitryand/or code that may be adapted to receive a delay signal and channelestimates ĥ₁ . . . ĥ_(N) and generate a matrix of baseband combinedchannel estimates Ĥ_(2×N) and an algorithm start signal to the SWGalgorithm block 320 (FIG. 3A). The phase rotator 502 may comprisesuitable circuitry, logic and/or code that may be adapted to receive adelay signal and generate an output signal to the complex combiner 506.The complex combiner 506 may be adapted to receive a plurality ofsignals from the phase rotator 502 and the channel estimate ĥ₁ andgenerate a phase rotated output of the channel estimate ĥ₁ to the firstintegrator 504. The first integrator 504 may comprise suitable logic,circuitry and/or code that may be adapted to receive at least a delaysignal and the output generated by the complex combiner 506. Based onthese received inputs, the first integrator 504 may generate a channelestimate between the first transmit antenna Tx_1 302 and the secondreceive antenna Rx_2 304, ĥ₂₁, to the SWG algorithm block 320 (FIG. 3A).The second integrator 508 may comprise suitable logic, circuitry and/orcode that may be adapted to receive at least a delay signal and thechannel estimate and generate a channel estimate between the firsttransmit antenna Tx_1 302 and the first receive antenna Rx_(—1) 302,ĥ₁₁, to the SWG algorithm block 320 (FIG. 3A). Similarly, the channelsbetween the remaining transmit antennas and the two receiving antennasRx_(—1) 302 and Rx_(—2) 304 may be estimated.

In operation, as an example, the channel estimator 500 may determine, atthe baseband, a combined estimate of the two baseband combined channelestimates between the two receive antennas and the first transmitantenna Tx_1 (FIG. 3A) as:ĥ ₁ =ĥ ₁₁ +e ^(jw) ^(r) ^(t) ĥ ₂₁,where w_(r)=2πf_(r) and f_(r) is the rotation frequency. A channelestimate of the first receive antenna (ĥ₁₁) may be determined by takingthe expected value or integration of ĥ₁ over a 0-360 degree rotation sothat:ĥ ₁₁ =E[ĥ ₁₁ +e ^(jw) ^(r) ^(t) ĥ ₂₁ ]=ĥ ₁₁ +E[e ^(jw) ^(r) ^(t) ĥ ₂₁],where E[e^(jw) ^(r) ^(t)ĥ₂₁] over a full rotation is equal to zero.Channel estimate of the second antenna (ĥ₂₁) may be determined by takingthe expected value or integration of ĥ₁ multiplied by a complexconjugate of the rotation waveform over a 0-360 degree rotation period.In this case, the channel estimate may be expressed as:ĥ ₂₁ =E[e ^(−jw) ^(r) ^(t) ĥ ₁ ]=E[e ^(−jw) ^(r) ^(t)(ĥ ₁₁ +e ^(jw) ^(r)^(t) ĥ ₂₁)]=E[e ^(−jw) ^(r) ^(t) ĥ ₁₁ +ĥ ₂₁ ]=E[e ^(−jw) ^(r) ^(t) ĥ ₁₁]+ĥ ₂₁,where E[e^(−jw) ^(r) ^(t)ĥ₁₁] over a full rotation is equal to zero.Similarly, channel estimates related to the combined channel estimatesĥ₂ . . . ĥ_(N) can be determined to obtain the matrix of channelestimates Ĥ_(2×N). Following this example, the matrix of channelestimates Ĥ_(K×N) of FIG. 3B may be determined similarly by integrationor multiplication by the complex conjugate of the rotation waveformse^(jw) ^(r1) ^(t) to e^(jw) ^(r(K−1)) ^(t) and integration. In addition,the above channel estimator operation example may be extended to Mreceive antenna system with N RF chains RF₁ . . . RF_(N) to form thepropagation channel estimate matrix Ĥ_(M×N) of FIG. 1. The actual timevarying channel impulse response estimates ĥ_(xy) may comprise multiplepropagation paths arriving at different delays. In that regard, thematrix Ĥ_(M×N) of the propagation channel estimates may consist ofmultiple path estimates arriving at different delays. For each patharriving at a different delay, channel matrix estimate Ĥ_(M×N) may bedetermined following the channel estimator 500 operation for each path.

The matrix Ĥ_(2×N) may be represented as:

${\hat{H}}_{2 \times N} = \begin{bmatrix}{\hat{h}}_{11} & {\hat{h}}_{12} & \ldots & {\hat{h}}_{1N} \\{\hat{h}}_{21} & {\hat{h}}_{22} & \ldots & {\hat{h}}_{2N}\end{bmatrix}$

Rotation on the additional antennas may be performed continuously, but apreferred embodiment is to perform the rotation periodically, as shownin FIG. 2. A continuous rotation may be perceived by the modem as a highDoppler, and for some modem implementations this may decrease the modemperformance. The period between consecutive rotations may depend on theDoppler frequency. At a higher Doppler frequency, it may be necessary tomore frequently track the channel, while at a lower Doppler frequency,tracking may be less frequent. The period may also depend on the desiredmodem performance and channel estimation accuracy. For example, if theDoppler frequency is 5 Hz, then a period between consecutive rotationsof 1/50 sec. may be chosen. This results in 10 rotations or channelestimations per signal fade. The time duration of the rotation itselfmay be selected based on the channel estimation accuracy andcorresponding modem performance. Generally, longer rotation time resultsin a better channel estimate because of the longer integration period.

The antenna rotation technique may be extended to multiple receiveantennas (K) belonging to a single RF chain, RF₁ of FIG. 1 for example,as shown in the wireless system in FIG. 3B. In that case, K−1 antennamultiplying waveforms may be used which are orthogonal to each other.

FIG. 6 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized forchannel estimation in a 2-Tx and M-Rx antennas system, in accordancewith an embodiment of the invention. Referring to FIG. 6, after startstep 602, in step 604, the phase rotator start controller 314 in FIG. 3Bmay receive the reset signal to initiate operations for determiningpropagation channel estimates and channel weights in the SWBBG 310. Thephase rotator start controller 314 may generate control signals to thedelay block 316 and to the RF phase and amplitude controller 312. Thecontrol signals to the delay block 316 may be utilized to determine adelay time to be applied by the delay block 316. The control signals tothe RF phase and amplitude controller 312 may be utilized to determinewhen to apply the rotation waveforms that have been modified by thechannel weights to the mixers 308 to 309.

In step 606, the RF phase and amplitude controller 312 may applyrotation waveforms, such as those provided by the rotation waveformsources 342 in FIG. 3C, to the mixers 308 to 309 in FIG. 3B. In step608, the delay block 316 may apply a time delay signal to the SWGchannel estimator 318 to reflect the interval of time that may occurbetween receiving the single channel communication signals and when thefirst and second baseband combined channel estimates, {circumflex over(h)}₁ and {circumflex over (h)}₂, are available to the SWG channelestimator 318. For example, the time delay signal may be utilized as anenable signal to the SWG channel estimator 318, where the assertion ofthe time delay signal initiates operations for determining propagationchannel estimates. In step 610, the SWG channel estimator 318 mayprocess the first and second baseband combined channel estimates,{circumflex over (h)}₁ and {circumflex over (h)}₂, and may determine thematrix Ĥ_(K×2) of propagation channel estimates ĥ₁₁ to ĥ_(K1) and ĥ₁₂ toĥ_(K2). The SWG channel estimator 318 may transfer the propagationchannel estimates ĥ₁₁ to ĥ_(K1) and ĥ₁₂ to ĥ_(K2) to the SWG algorithmblock 320. In step 612, the SWG channel estimator 318 may generate thealgorithm start signal and may assert the signal to indicate to the SWGalgorithm block 320 that it may initiate operations for determiningchannel weights.

In step 614, the SWG algorithm block 320 may determine the channelweights comprising phase and amplitude components, A_(K−1) and φ₁ toφ_(K−1), based on the propagation channel estimates ĥ₁₁ to ĥ_(K1) andĥ₁₂ to ĥ_(K2) and/or noise power estimates, for example. The SWGalgorithm block 320 may transfer the channel weights to the RF phase andamplitude controller 312. In some instances, the SWG algorithm block 320may also generate the weight factors W₁ and/or W₂. In step 616, the SWGalgorithm block 320 may generate the algorithm end signal to indicate tothe RF phase and amplitude controller 312 that the channel weights areavailable to be applied to the mixers 308 to 309. In step 618, the RFphase and amplitude controller 312 may apply the rotation waveforms withphase and amplitude components, A_(K−1) and φ₁ to φ_(K−1), to the mixers308 to 309, in accordance with the control signals provided by the phaserotator start controller 314.

In step 620, the receiver system 330 in FIG. 3B may determine whetherthe phase rotation operation on the received single channelcommunication signals is periodic. When the phase rotation operation isnot periodic but continuous, the process may proceed to step 608 where adelay may be applied to the SWG channel estimator 318. In instances whenthe phase rotation operation is periodic, the process may proceed tostep 622 where the receiver system 330 may wait until the next phaserotation operation is initiated by the reset signal. In this regard, theprocess control may proceed to step 604 upon assertion of the resetsignal to the phase rotator start controller 314.

FIG. 7 is a flowchart illustrating exemplary steps that may be utilizedfor channel estimation in a wireless communication system, in accordancewith an embodiment of the invention. Referring to FIG. 7, the exemplarysteps may start at step 702. In step 704, a plurality of communicationsignals may be received by a plurality of receive antennas. In step 706,the phase at the additional antennas may be rotated. In step 710, a lowpass filter may filter the received plurality of communication signals.In step 712, a low noise amplifier may amplify the received plurality ofcommunication signals. In step 714, an analog to digital converter mayconvert the received plurality of communication signals into digitalsignals. In step 716, a chip matched filter may modulate the receivedplurality of communication signals into in phase (I) and quadrature (Q)components.

In step 718, a plurality of vectors of baseband combined channelestimates ĥ₁ . . . ĥ_(N) may be generated by a baseband processor. Instep 720, a Ĥ_(M×N) of processed baseband combined channel estimates maybe generated based on receiving the generated plurality of vectors ofbaseband combined channel estimates. In step 724, a weight generator maydetermine a plurality of weights that may be applied to each of thereceived plurality of communication signals to maximize the receiversignal-to interference-to-noise-ratio (SINR), for example. In step 726,the plurality of weights may be calculated by utilizing an adaptivealgorithm, for example, a least mean squares algorithm. In step 728, thebaseband processor may be adapted to adjust a phase and an amplitude ofat least a portion of the received plurality of communication signalsbased on the generated plurality of amplitude and phase correctionsignals A_(i) and φ_(i) respectively and the generated plurality ofamplitude and phase correction signals A_(i) and φ_(i) respectively maybe applied to the mixers at receiving antennas. Control then passes tostep 606 for periodic channel estimation and amplitude and phasecorrection.

Another embodiment of the invention may provide a machine-readablestorage, having stored thereon, a computer program having at least onecode section executable by a machine, thereby causing the machine toperform the steps as described above for channel estimation in amulti-input multi-output (MIMO) system.

In another embodiment of the invention a system for channel estimationin a communication system may be provided. With reference to FIG. 1E, aplurality of receive antennas 108 _(1 . . . M) may be adapted to receivea plurality of communication signals from a plurality of transmitantennas 106 _(1 . . . N). A baseband processor BB 118 may be adapted togenerate a plurality of baseband combined channel estimates {circumflexover (h)}₁ to {circumflex over (h)}_(P) and a plurality of estimates{circumflex over (X)}₁ to {circumflex over (X)}_(P) of the originalinput signals X₁ to X_(P) based on phase rotation in response toreceiving the plurality of communication signals RF_(1 . . . P). Thesingle weight baseband processor SWGBB 121 may generate a matrix ofprocessed baseband combined channel estimates Ĥ based on receiving thegenerated plurality of vectors of baseband combined channel estimates{circumflex over (h)}₁ to {circumflex over (h)}_(P).

The single weight baseband processor SWGBB 121 may generate a pluralityof amplitude and phase correction signals A_(i) and φ_(i) based on thegenerated plurality of vectors of baseband combined channel estimates{circumflex over (h)}₁ to {circumflex over (h)}_(P). The SWG 110 mayutilize the generated plurality of amplitude and phase correctionsignals to modify the phase and amplitude of at least a portion of thetransmitted signals received by the plurality of receive antennas 108_(1 . . . M) and generate a plurality of output signals RF_(1 . . . P).The single weight baseband processor SWGBB 121 may be adapted to adjusta phase and an amplitude of at least a portion of the received pluralityof communication signals based on the generated plurality of amplitudeand phase correction signals A_(i) and φ_(i) respectively.

A weight generator SWG 110 may determine a plurality of weights that maybe applied to each of the received plurality of communication signalsbased on the generated plurality of amplitude and phase correctionsignals. The single weight baseband processor SWGBB 121 may be adaptedto adjust a phase and an amplitude of at least a portion of the receivedplurality of communication signals based on the determined plurality ofweights. The baseband processor SWGBB 121 may be adapted to calculatethe determined plurality of weights by utilizing an adaptive algorithm.The adaptive algorithm may be a least mean squares algorithm, forexample. A low noise amplifier may be adapted to amplify the receivedplurality of communication signals. The cluster path processors 116_(1 . . . P) may be adapted to generate the plurality of vectors ofbaseband combined channel estimates {circumflex over (h)}₁ to{circumflex over (h)}_(P) via rotation at additional antennas. Thesystem may comprise circuitry that may be adapted to modulate thereceived plurality of communication signals into in phase (I) componentsand quadrature (Q) components.

Accordingly, the present invention may be realized in hardware,software, or a combination of hardware and software. The presentinvention may be realized in a centralized fashion in at least onecomputer system, or in a distributed fashion where different elementsare spread across several interconnected computer systems. Any kind ofcomputer system or other apparatus adapted for carrying out the methodsdescribed herein is suited. A typical combination of hardware andsoftware may be a general-purpose computer system with a computerprogram that, when being loaded and executed, controls the computersystem such that it carries out the methods described herein.

The present invention may also be embedded in a computer programproduct, which comprises all the features enabling the implementation ofthe methods described herein, and which when loaded in a computer systemis able to carry out these methods. Computer program in the presentcontext means any expression, in any language, code or notation, of aset of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiment disclosed, but that the present invention willinclude all embodiments falling within the scope of the appended claims.

What is claimed is:
 1. A method for channel estimation in communicationsystem, the method comprising: generating a plurality of vectors ofbaseband combined channel estimates based on phase rotation of areceived plurality of communication signals, said phase rotation beingdetermined based on orthogonalization of channels associated with saidreceived plurality of communication signals; generating a plurality ofamplitude correction signals based on said generated plurality ofvectors of baseband combined channel estimates; and adjusting anamplitude of a portion of said received plurality of communicationsignals based on a portion of said generated plurality of amplitudecorrection signals, wherein at least one step of said method isperformed by at least one hardware device.
 2. The method according toclaim 1, further comprising; receiving said plurality of communicationsignals from a plurality of transmit antennas.
 3. The method accordingto claim 1, farther comprising: generating a matrix of processedbaseband combined channel estimates based on said generated plurality ofvectors of baseband combined channel estimates.
 4. The method accordingto claim 1, further comprising: determining a plurality of weights to beapplied to each of said received plurality of communication signalsbased on said generated plurality of amplitude correction signals. 5.The method according to claim 4, wherein the step of adjustingcomprises: adjusting said amplitude based on said determined pluralityof weights.
 6. The method according to claim 4, further comprising:calculating said determined plurality of weights by utilizing anadaptive algorithm.
 7. The method according to claim 6, wherein the stepof adjusting comprises: calculating said determined plurality of weightsusing a least mean squares algorithm.
 8. The method according to claim1, wherein the step of generating said plurality of vectors of basebandcombined channel estimates comprises: generating said plurality ofvectors of baseband combined channel estimates via rotation atadditional antennas.
 9. The method according to claim 1, furthercomprising: modulating said received plurality of communication signalsinto in phase (I) components and quadrature (Q) components.
 10. A methodfor channel estimation in a communication system, comprising: generatinga plurality of vectors of baseband combined channel estimates based onphase rotation of a received plurality of communication signals, saidphase rotation being determined based on orthogonalization of channels,associated with said received plurality of communication signals;generating a plurality of phase correction signals based on saidgenerated plurality of vectors of baseband combined channel estimates;and adjusting a phase of a portion of said received plurality ofcommunication signals based on a portion of said generated plurality ofphase correction signals, wherein at least one step of said method isperformed by at least one hardware device.
 11. A system for channelestimation in a communication system, comprising: one or more circuitsfor use in a receiver, said one or more circuits being configured to:generate a plurality of vectors of baseband combined channel estimatesbased on phase rotation of a received plurality of communicationsignals, said phase rotation being determined based on orthogonalizationof channels associated with said received plurality of communicationsignals; generate a plurality of amplitude correction signals based onsaid generated plurality of vectors of baseband combined channelestimates; and adjust an amplitude of a portion of said receivedplurality of communication signals based on a portion of said generatedplurality of amplitude correction signals.
 12. The system according toclaim 11, wherein said one or more circuits are further configured toreceive said plurality of communication signals from a plurality oftransmit antennas.
 13. The system according to claim 11, wherein saidone or more circuits are further configured to generate a matrix ofprocessed baseband combined channel estimates based on said generatedplurality of vectors of baseband combined channel estimates.
 14. Thesystem according to claim 11, wherein said one or more circuits arefurther configured to determine a plurality of weights to be applied toeach of said received plurality of communication signals based on saidgenerated plurality of amplitude correction signals.
 15. The systemaccording to claim 14, wherein said one or more circuits are configuredto adjust said amplitude based on said determined plurality of weights.16. The system according to claim 14, wherein said one or more circuitsare further configured to calculate said determined plurality of weightsby utilizing an adaptive algorithm.
 17. The system according to claim16, wherein said adaptive algorithm is a least mean squares algorithm.18. The system according to claim 11, wherein said one or more circuitsare configured to generate said plurality of vectors of basebandcombined channel estimates via rotation at additional antennas.
 19. Thesystem according to claim 11, wherein said one or more circuits arefurther configured to modulate said received plurality of communicationsignals into in phase (I) components and quadrature (Q) components. 20.The system according to claim 11, wherein said one or more circuits arefurther configured to adjust a phase of said portion of said receivedplurality of communication signals based on a portion of a generatedplurality of phase correction signals, wherein said generated pluralityof phase control signals are generated based on said generated pluralityof vectors of baseband combined channel estimates.